This invention is in the field of digital communications, and is more specifically directed to the operation of digital subscriber line communications sessions.
Digital Subscriber Line (DSL) technology has become a primary technology providing high-speed Internet access, and now video and telephone communications, in the United States and around the world. As is well known in the art, DSL communications are carried out over existing telephone “wire” facilities, between individual subscribers and a central office (CO) location, operated by a telephone company or an Internet service provider. Typically, some if not all of the length of the loop between the CO and the customer premises equipment (CPE) consists of conventional twisted-pair copper telephone wire.
Modern DSL communications use multicarrier modulation (MCM) techniques, more specifically discrete multitone modulation (DMT), by way of which the data signals are modulated onto orthogonal tones, or subcarriers. within a relatively wide frequency band (on the order of 1.1 MHz for conventional ADSL, and on the order of 2.2 MHz for ADSL2+), residing above the telephone voice band. The data symbols modulated onto each subchannel are encoded as points in a complex plane, according to a quadrature amplitude modulation (QAM) constellation. The number of bits of data that are carried over each subchannel (i.e., the “bit loading”), and thus the number of points in the QAM constellation for that subchannel, depend on the signal-to-noise ratio (SNR) at the subchannel frequency, which in turn depends on the noise and signal attenuation present at that frequency. For example, subchannels with low noise and low attenuation may carry data in ten-bit to fifteen-bit symbols, represented by a relatively dense QAM constellation with short distances between constellation points. On the other hand, noisy channels may be limited to only two or three bits per symbol, requiring a greater distance between adjacent points in the QAM constellation to resolve the transmitted symbol. The sum of the bit loadings over all of the subchannels in the transmission band for a DSL link of course amounts to the number of transmitted bits per DSL symbol for that link. And the data rate for DSL communications corresponds to the product of the symbol rate with the number of bits per DSL symbol.
FIG. 1 illustrates the data flow in conventional DSL communications, in a single direction (e.g., downstream, from a central office “CO” to customer premises equipment “CPE”). Typically, each DSL modem (i.e., both at the CO and also in the CPE) includes a transceiver (i.e., both a transmitter function and a receiver function), so that data is also communicated in the opposite direction over transmission channel LP according to a similar DMT process. As shown in FIG. 1, the input bitstream that is to be transmitted, typically a serial stream of binary digits in the format as produced by the data source, is applied to constellation encoder 11 in a transmitting modem 10. Constellation encoder 11 fundamentally groups the bits in the input bitstream into multiple-bit symbols that are used to modulate the DMT subchannels, with the number of bits in each symbol determined according to the bit loading assigned to its corresponding subchannel, based on the characteristics of the transmission channel as mentioned above. Encoder 11 may also include other encoding functions, such as Reed-Solomon or other forward error correction coding, trellis coding, turbo or Low Density Parity Check Codes (LDPC) coding, and the like. The symbols generated by constellation encoder 11 correspond to points in the appropriate modulation constellation (e.g., QAM), with each symbol associated with one of the DMT subchannels. Following constellation encoder 11, shaping function 12 derives a clip prevention signal included in the encoded signals to be modulated, to reduce the peak-to-average ratio (PAR) as transmitted as described in commonly assigned U.S. Pat. No. 6,954,505, issued Oct. 11, 2005, and incorporated herein by this reference.
These encoded symbols are applied to inverse Discrete Fourier Transform (IDFT) function 13, which associates each symbol with one subchannel in the transmission frequency band, and generates a corresponding number of time domain symbol samples according to the Fourier transform. As known in the art, cyclic insertion function 14 appends a cyclic prefix or suffix, or both, to the modulated time-domain samples from IDFT function 13, and presents the extended block of serial samples to parallel-to-serial converter 15. Cyclic insertion function 14 may follow rather than precede parallel-to-serial converter 15 in the transmission sequence, in some implementations. In either case, the time-domain serial sequence, as may be upsampled (not shown) as appropriate, is applied to digital filter function 16, which processes the datastream in the conventional manner to remove image components and voice band or Integrated Services Digital Network (ISDN) interference. The filtered digital datastream signal is converted into the analog domain by digital-to-analog converter 17. After conventional analog filtering and amplification (not shown), the resulting DMT signal is transmitted over a channel LP, over some length of conventional twisted-pair wires, to a receiving DSL modem 20, which, in general, reverses the processes performed by the transmitting modem to recover the input bitstream as the transmitted communication.
At receiving DSL modem 20, analog-to-digital conversion 22 converts the filtered analog signal into the digital domain, following which conventional digital filtering function 23 is applied to augment the function of pre-conversion receiver analog filters (not shown). A time domain equalizer (TEQ) (not shown) may apply a finite impulse response (FIR) digital filter to effectively shorten the length of the impulse response of the transmission channel LP. After removal of the cyclic extension from each received block as performed by function 25, serial-to-parallel converter 24 converts the datastream into a number of samples (2N) for application to Discrete Fourier Transform (DFT) function 27. DFT function 27 recovers the modulating symbols at each of the subchannel frequencies, by reversing the IDFT performed by function 13 in transmission. The output of DFT function 27 is a frequency domain representation of the transmitted symbols multiplied by the frequency-domain response of the effective transmission channel. Frequency-domain equalization (FEQ) function 28 divides out the frequency-domain response of the effective channel, recovering the modulating symbols, each representable as a point in a QAM constellation. Constellation decoder function 29 then resequences the symbols into a serial bitstream, decoding any encoding that was applied in the transmission of the signal and producing an output bitstream that corresponds to the input bitstream upon which the transmission was based. This output bitstream is then forwarded to the client workstation, or to the central office network, as appropriate for the location.
In practice, as mentioned above, the data rates achievable for a given DSL loop are generally limited by the level of noise in transmission channel LP. Such noise is typically dominated by crosstalk, both crosstalk in the received signal that results from transmissions in the reverse direction over the same facility (i.e., “near-end” crosstalk, or “NEXT”), and also crosstalk from communications traveling in the same direction in other conductors within the same physical binder (i.e., “far-end” crosstalk, or “FEXT”). Noise not associated with DSL communications, such as narrowband interference from a nearby RF interferer, also adversely affects the achievable data rate.
A measure of the robustness of a transmission system is its “margin”. In DSL communications, the margin is typically defined as the amount of additional signal-to-noise margin, at a given data rate, that is beyond the minimum signal-to-noise ratio required for the given line code (e.g., the QAM code). Typical line codes used in DSL communications are characterized by a signal-to-noise ratio gap (i.e., commonly referred to simply as the “gap”) Γ, which is a function of a selected probability of symbol error; the gap Γ is effectively a measure of the efficiency of the transmission method on an additive white Gaussian noise channel. According to Shannon's information theory, one can quantify the achievable bit rate b of a line code, at a given error probability and a specified bandwidth BW, using the gap Γ as follows:
      b    _    =      BW    ·                  log        2            ⁡              (                  1          +                      SNR            Γ                          )            for a given signal-to-noise ratio SNR. Those skilled in the art will recognize that, from this expression, the bit loading bi for a QAM modulated subchannel i is derived as:
      b    i    =            log      2        ⁡          (              1        +                  SNR          Γ                    )      Conversely, one may derive the margin γ as:
  γ  =      SNR          Γ      ⁡              (                              2            b                    -          1                )            for a current value b of the bits per DSL symbol. This margin is, in effect, the ratio of the current signal-to-noise ratio SNR with the minimum signal-to-noise ratio for the current bit rate b based on the gap Γ for the line code being used.
Accordingly, the bit rate (i.e., bits per DSL symbol) of a DSL communications session determines the margin, and thus the ability of the session to tolerate additional noise, or an additional disturbance, or a sudden change in the loop condition. A DSL communications session that is operating at its maximum achievable bit rate b cannot tolerate a substantial increase in noise or any other substantial degradation in loop conditions.
Several approaches for changing the bit loadings of an established DSL communications session, for example in light of changes in the channel or operating conditions, are known in the art. According to one approach, referred to as “bit swapping”, the number of bits allocated to one or more subchannels is modified during “showtime” (i.e., during live communications of payload data) without interrupting the communications session or requiring retraining of the transceivers. An example of the bit swapping technique is described in commonly assigned U.S. Pat. No. 5,400,322, issued Mar. 21, 1995, and incorporated herein by reference. Bit swapping is also described at §11.2 of the ADSL standard ITU-T G.992.1, “Asymmetric digital subscriber line (ADSL) transceivers” (International Telecommunication Union, Jun. 1999), such standard incorporated herein by reference. In practice, the bits that are removed from the bit loading of one subchannel are added to another subchannel, so that the number of bits per DSL symbol remains constant. As such, the overall data rate is not modified by the bit swapping technique, and either the CO or the CPE modem can initiate the bit swap.
Another technique, referred to as “Seamless Rate Adaptation”, or “SRA”, permits changes in the bit loadings and data rates, without necessarily changing the parameters involved in forward error correction coding, interleaving, or framing. SRA thus enables changes in the bit loadings and bit rate without interrupting the data flow (i.e., “seamlessly”). According to the SRA technique, the modulation layer is functionally decoupled from the framing layer, so that changes in the bit loadings and the data rate do not affect the framing parameters. Accordingly, SRA modifications do not cause loss of frame synchronization, and thus do not necessarily interrupt data flow. SRA is described in §10.2 of the ADSL standard ITU-T G.992.3, “Asymmetric digital subscriber line transceivers 2 (ADSL2)”, (International Telecommunication Union, Jan. 2005), such standard incorporated herein by reference.
By way of further background, the ADSL2 and ADSL2+ standards define certain low power, or quiescent, operating modes for DSL modems. These operating modes are indicated by specific states of the DSL link. The “L2” low power state is a low traffic mode, in which downstream traffic is limited to only background data. Upstream traffic is not affected in the L2 state. As such, the L2 state enables substantial power savings in the CO modem, but little or no power savings for CPE equipment. The “L3” state is a “no traffic” mode, in which both the CO and CPE modems idle and in a very low power condition, with no signal transmitted either in the downstream or upstream directions. The normal operating mode, in which the ADSL link is fully functional, with both upstream and downstream traffic enabled, is referred to as the “L0” state. A description of power management operations involving the L0, L2, and L3 states is provided in §9.5 of the ADSL2 standard ITU-T G.992.3, incorporated by reference above.
As known in the art, and as described in the ADSL2 standard, the CO modem requests entry into the L2 low power state, from the L0 “full-on” state, for example in response to detecting low traffic in the downstream direction over the link over a defined period of time. Upon the CPE modem granting the L2 request, the CO modem can then enter the L2 state. Transition from the L0 state to the L3 “idle” state can be requested by either the CO modem or the CPE modem, for example upon either modem detecting the absence of traffic in either direction for a given time. In response to granting of the L3 state by the non-requesting transceiver, then an orderly shutdown procedure is performed at each side of the DSL link; the CO modem can also effect a “disorderly” shutdown of the link to enter the L3 state, for example if power is suddenly lost at the CPE modem.
The transition from the L2 low power state to the L0 full-on state can be performed quite rapidly at the CO modem. An exit from the L2 state can be requested by either the CO modem or by the CPE modem, in response to which the CO modem returns the link to the L0 state by way of a specified fast exit sequence. Transition from the L3 idle state to the L0 full power state requires an initialization sequence, according to the current standards.
In the ADSL2 and ADSL2+ standards, a “short” initialization sequence is provided for the transition from the L3 state to the L0 state, and also as a fast recovery from significant line condition changes occurring during showtime. This short initialization (also referred to in the art as “fast initialization”) is described at §8.14 of the G.992.3 standard, incorporated by reference above. According to this short initialization procedure, certain operations in the full training and initialization procedure are skipped entirely (e.g., the so-called “G.994.1” phase, in which handshaking and the setting of power and spectral shaping parameters occur), and certain other phases are performed over limited durations, as compared to full initialization.
These low power modes and the corresponding short initialization procedure that are permitted under ADSL2 have become especially important considering the many new applications enabled by DSL technology. In practice, many DSL subscribers simply leave their DSL modems, as well as their computers, in an “always on” condition; the L3 idle state thus can save substantial energy over time, and especially as accumulated over a large number of CPE modems. In addition, many subscribers now also use Voice over Internet Protocol (VoIP) for their land-line telephone service, in which case the subscribers make and receive telephone calls via their DSL modems. As such, the initialization process to make a transition from the L3 idle state at the CPE modem into its full-on L0 state must be kept short, for example on the order of one second or less. Any further delay in re-activation of the DSL link becomes especially noticeable to the user, especially if this delay appears to the user in waiting for a dial tone, waiting for the DSL modem to initialize upon answering a call, or in waiting for a receiving VoIP subscriber to initialize his or her CPE equipment upon receiving a phone call from the user.
One of the operations performed in short initialization, over a limited duration, is channel analysis, which determines the characteristics of the transmission channel over the subchannels of the transmission band. As a result of that channel analysis, if the transmission parameters from the previous L0 state for the link are still adequate for the current condition of the line determined in this limited channel analysis, those previous parameters may be recalled from memory at both the CO modem and the CPE modem, and need not be re-communicated in the “exchange” phase of the short initialization procedure. As known in the art, these parameters include the frame multiplexor control parameters Lp that specify the number of bits communicated for each latency path p, in each transmission direction, of each data frame, as defined in the G.992.3 standard incorporated above. The total data rate is thus expressed as ΣLp. The parameters to be exchanged in the short initialization process, if necessary, also include the “bits and gains” for each of the subchannels in the transmission bandwidth, in each direction; the “bits” bi of course specify the bit loading for subchannel i, and the “gains” gi specify the scale factor to be applied to that subchannel i, relative to the nominal gain for that subcarrier as used in channel analysis.
It has been observed, according to this invention, that the time required to exchange these parameters Lp, bi, gi in this short initialization procedure can be as much as 0.3 seconds. This time is dominated by the exchange of the parameters from the CPE modem to the CO modem (i.e., the so-called R-PARAMS phase), considering that the CPE modem will be communicating the parameters to be applied by the CO for the large number of downstream subchannels (e.g., as many as 512 subchannels), but communicating these subchannels only over the much smaller (e.g., thirty-two) number of subchannels allocated to upstream communications. It is contemplated, according to this invention, that the addition of these 0.3 seconds for parameter exchange to the approximately one second delay required for short initialization will be readily noticeable to the users, especially in making and answering VoIP telephone calls.